1. Field of the Invention
The present invention relates generally to a control system for controlling revolution speed of an electric motor, such as an induction motor. More specifically, the invention relates to a motor speed control system with improved precision in speed control by compensating error factors which may otherwise cause error is motor speed control.
2. Description of the Background Art
As is well known, variable speed electric motors have been applied in various facilities, such as an elevator car drive system and so forth. In case that the electric motor, such as an induction motor is employed for an elevator car drive system, revolution speed of the electric motor has to be controlled according to a preset known schedule including acceleration stage and deceleration stage. In order to control the revolution speed of the electric motor, various control systems have been employed. The control system typically comprises an electric power converter and inverter for driving the electric motor at a controlled speed. The converter typically comprises a direct current (DC) rectifier for rectifying three-phase alternating current (AC) power and for supplying the rectified DC power to the inverter. The inverter comprises a plurality of pairs of series-connected switching elements to generate an adjustable frequency output. In many applications, frequency adjustment is effected through a control circuit which employs a pulse width modulated (PWM) control technologies for producing variable frequency gate output to periodically switch the motor at various speed. In the practical control, the electric motor is driven in motoring mode in the acceleration stage of the elevator car driving schedule for increasing the motor speed, and in braking mode in the deceleration stage of the elevator car driving schedule for decreasing the motor speed.
For adjusting the inverter output frequency and amplitude, various control technologies have been employed, such as proportional/integral (PI) control, vector control and so forth. In general, the motor speed control has been performed employing FEEDBACK or CLOSED LOOP control technologies for adjusting the motor revolution speed to a desired speed which is determined according to the preset motor drive schedule. For this, the actual revolution speed of the electric motor is monitored and compared with the desired speed to produce a speed error signal. Based on this speed error signal, PI control is performed for reducing the error to zero. For performing motor speed control employing the PI control technologies, a PI control circuit is provided. The PI control circuit has a transfer function Gc(S) which can be illustrated by the following equation: EQU Gc(S)=Kp+Ki/S
where Kp and Ki are control constants.
The PI control circuit receives the actual motor speed data Wc as FEEDBACK data and compares the same with the desired speed data Ws in order to derive a torque control signal Itc. The torque control signal Itc is applied to the electric motor for adjusting the revolution speed for decreasing the error between the actual motor speed Wc and the desired speed Ws toward zero (0).
In PI control technology, the response characteristics are determined by the control constants Kp and Ki. Generally, these control constants Kp and Ki are set in consideration of the inertia moment of the load on the electric motor. Conventionally, the control constants Kp and Ki are set as fixed values. In case of the elevator car speed control, the load of the electric motro varies significantly depending upon the passengers in the elevator car. Since variation magnitude of the load on the electric motor is substantial, a difficulty has been encountered in obtaining optimal response characteristics at any load condition.
In the alternative, vector control technologies have also been employed in the motor speed control. The basic idea of the vector control is to divide a primary current into an excitation current and a secondary current, in order to control each independently of each other. The flux of the excitation current and the vector of the secondary current are so established as to cross perpendicularly to each other. For improving response characteristics and precision in vector control, interference between the secondary flux component and secondary current component is avoided. The interference avoidance technology has been disclosed in Japanese Patent First (unexamined) Publication 59-165982, for example.
In such vector control, basic electric equation illustrating the induction motor by two axes d-q rotating at electric angular velocity .omega. is as follows: ##EQU1## On the other hand, the torque T can be illustrated by the following equation: EQU T=K.times.(i.sub.2 d.times.i.sub.1 q-i.sub.2 q.times.i.sub.1 d)(2)
where
V.sub.1 d, V.sub.1 q are primary voltage of d axis and q axis; PA1 i.sub.1 d, l.sub.1 q are primary current of d axis and q axis; PA1 i.sub.2 d, l.sub.2 q are secondary current of d azis and q axis; PA1 R.sub.1, R.sub.2 are primary and secondary resistances; PA1 L.sub.1, L.sub.2 are primary and secondary inductances; PA1 M is relative inductance of primary and secondary inductances; PA1 P is d/dt PA1 .omega..sub.s is slip frequency; and PA1 K is constant. PA1 I.sub.o is a set excitation current value; PA1 I.sub.T is a set torque current value; PA1 I.sub.o ' is a core loss current value; PA1 R.sub.2 is a secondary resistance; PA1 S is a slip; PA1 rm is a core loss resistance; PA1 L.sub.2 is a secondary inductance; and PA1 M is a relative inductance PA1 B=(1+rm.sup.2 /.omega..sup.2 M.sup.2).times.(M.sup.2 /L.sub.2).times.I.sub.o PA1 I.sub.T * is torque current command; PA1 I.sub.o * is excitation current command; PA1 M is a relative inductance; PA1 L.sub.2 is a secondary inductance; PA1 R.sub.2 is a secondary resistance; and PA1 .omega. is a power source angular frequency.
In the equation set forth above, the q axis is set as the axis of the secondary current and d axis is set as the axis of the flux. Mutual interference of the secondary flux and the secondary current is compensated in order to derive the primary voltages V.sub.1 d and V.sub.1 q based on the excitation current command i.sub.O * and torque current command i.sub.T *. In the practical vector control, angular velocity .omega..sub.r of a rotor in the electric motor is monitored by means of a pick-up. The monitored angular velocity .omega..sub.r is compared with the speed command .omega.* to derive the torque current command i.sub.T *. Based on this torque current command i.sub.T *, the excitation current command i.sub.O *, and secondary time constant .tau..sub.2, slip frequency .omega..sub.s is derived by the following equation: EQU .omega..sub.s =i.sub.T */(.tau..sub.2 .times.i.sub.O *)
Then a power source angular frequency .omega..sub.0 is derived by adding the slip frequency .omega..sub.s derived as above to the angular velocity .omega..sub.r of the rotor of the motor. Then, sin wave signal SIN .omega..sub.O and cos wave signal COS .omega..sub.O t respectively having power source frequency .omega..sub.O are produced. Utilizing the sin wave signal SIN .omega.Ot and the cos wave signal COS .omega.Ot, the voltage signal V.sub.1 d and V.sub.1 q for the axes d-q is derived on the basis of the excitation current command i.sub.O * and the torque current command i.sub.T *. These primary voltages V.sub.1 d and V.sub.1 q are 2/3 phase converted to generate three phase voltages ea*, eb* and ec*. PWM control for the inverter is thus performed by the three phase voltages ea*, eb* and ec* and a triangular wave in a known manner.
Such vector control is generally successful to improve the response characteristics in satisfactory level. However, due to presence of core loss, which is included in the torque current vector, the primary current in the braking mode becomes excessively smaller than normal value in relation to the primary current in the motoring mode. This clearly affects precision in motor speed control.